FIG. 11 is a block diagram showing an optical receiver employing a prior art transimpedance type preamplifier. In the FIG., a receiver 10 comprises a transimpedance type preamplifier 1 and a photodiode 5. A p-side electrode of the photodiode 5 is connected to an input of the transimpedance type preamplifier 1. The transimpedance type preamplifier 1 comprises a voltage amplifier 2 and a feedback resistor 3 which is attached to the voltage amplifier 2. That is, the feedback resistor 3 is connected in parallel between an input and output of the voltage amplifier 2 so as to feedback a part of the output signal of the voltage amplifier 2 to the input side. Reference numeral 4 designates an input terminal and numeral 6 designates an output terminal. Reference character P.sub.out shows light, reference character I.sub.in shows a current signal from the photodiode 5 and reference character V.sub.out shows a voltage signal output from the preamplifier 1.
A description is given of the operations.
Light P.sub.out entering the photodiode 5 is converted into a current signal I.sub.in through the photodiode 5 and enters into the preamplifier 1 from the input terminal 4. The current signal I.sub.in flows into the feedback resistor 3 if the input impedance of the voltage amplifier 2 is sufficiently high. In this case, when a gain of the voltage amplifier 2 is sufficiently high, suppose that the resistance value of the feedback resistor 3 is Rf[.OMEGA.] and the current signal amplitude is .DELTA.I.sub.in, then the output voltage amplitude .DELTA.V.sub.out is represented as follows: EQU .DELTA.V.sub.out =.DELTA.I.sub.in .multidot.Rf .multidot. (1)
In addition, the noise of the optical receiver 10 comprising the photodiode 5 and the preamplifier 1 is represented by the noise converted into input noise current density (unit:[A/ .sqroot.Hz]). This noise is controlled by a thermal noise due to the feedback resistor 3 and the channel noise of a source grounded field effect transistor (hereinafter referred to as an FET) in the voltage amplifier 2. The thermal noise &lt;i.sub.Rf 2&gt; (unit:[A/.sqroot.Hz]) due to the feedback resistor 3 is represented as follows: ##EQU1## where T is absolute temperature, K is Boltzmann's constant, and .DELTA. f is the operational bandwidth.
On the other hand, the 3dB bandwidth of the optical receiver 10 (f.sub.3dB) is, supposing that the capacitance of the photodiode 5 is C.sub.pd, the input capacitance of the preamplifier 1 is C.sub.in and the voltage gain of the voltage amplifier 2 is A, represented as follows: ##EQU2##
According to these formulae, in order to improve the sensitivity of the optical receiver 10 comprising the photodiode 5 and the preamplifier 1, it is effective to increase the feedback resistance value Rf of the feedback resistor 3 in the preamplifier 1 while maintaining the necessary bandwidth.
FIG. 12 shows the characteristic of the output voltage amplitude against the input current amplitude of the preamplifier 1. In the FIG., a curve obtained by connecting black dot .cndot. points shows the output voltage amplitude versus the input current amplitude when the feedback resistance value Rf is 2.2k1/3 at a transmission speed of 622 Mb/s and a curve obtained by connecting crossed marks.times.shows the output voltage amplitude against the input current amplitude when the feedback resistance value Rf is 2001/3 at a transmission speed of 622 Mb/s. FIGS. 13 to 16 are diagrams showing output voltage waveform versus input current amplitude when the feedback resistance value Rf is 2.2k1/3 and the input current amplitude is 0.2 mAp-p, 0.5 mAp-p, 1 mAp-p or 2 mAp-p, respectively. From these FIGS., it is apparent that, although it is, as described above, effective to increase the feedback resistance value Rf of the feedback resistor 3 in the preamplifier 1 to increase the sensitivity of the preamplifier 1, the dynamic input range is reduced due to an increase in the input impedance and the duty ratio of the output voltage signal unfavorably changes. More particularly, the output voltage signal is unfavorably distorted when a large signal, namely an input current having a large amplitude, is input as particularly shown in FIGS. 15 and 16.
Disclosed in Japanese Published Patent Application No. 3-54905 is a preamplifier in which the substantive feedback resistance value is changed between a value for inputting a large signal and a valve for inputting a small signal so as to enlarge the input dynamic range, thereby obtaining an output voltage signal without distortion when a large signal is input. FIG. 17 is a block diagram showing an optical receiver in which a diode is connected in parallel with a feedback resistor constituting the preamplifier similar to the optical receiver disclosed in Japanese Published Patent Application No. 3-54905. In the FIG., the same reference numerals as those of FIG. 11 designate the same or corresponding parts. A receiver 20 comprises a preamplifier 1a and the photodiode 5. In the preamplifier la, a diode 7 is connected in parallel with the feedback resistor 3 so that an anode of the diode 7 is connected to an input terminal 4 and a cathode thereof to an output terminal 6.
Next, a description is given of the operation.
Almost all of the current signal I.sub.in from the photodiode 5 flows into the feedback resistor 3 similarly as the optical receiver 10 shown in FIG. 11, and when the voltage drop of the feedback resistor 3 is less than the thresh-old voltage of a forward direction current flow of the diode 7 connected in parallel, a current almost flows into the feedback resistor 3 and the operation is the same as in the case with no diode 7. When the voltage drop of the feedback resistor 3 exceeds the threshold voltage of the forward direction current, a current also begins to flow through the diode 7 suppose that a resistance value when the diode 7 is "on" state is Rdf, the actual feedback resistance value is represented as follows: ##EQU3## Particularly, when Rf&gt;&gt;Rdf, the feedback resistance value Rfs is reduced to a value represented by the following formula (5) and the dynamic range is enlarged by that degree. EQU Rfs=Rdf (5)
FIG. 18 shows a characteristic of the output voltage amplitude versus the input current amplitude of the preamplifier la when the feedback resistance value Rf of the feedback resistor 3 is 2.2k1/3 and an internal resistance Rdf of the diode 7 in the "on" state is fixed at 1801/3 at a transmission speed of 622 Mb/s. The threshold voltage of the forward direction current (Schottky barrier voltage) of the diode 7 is dependent on a material of the diode 7. Here, a Schottky diode formed on a GaAs substrate is employed and the threshold up voltage of the forward direction current (Schottky barrier voltage) is approximately 0.6 V.
As is apparent from the FIG., when the input current amplitude to the preamplifier 1a is within 0.0 to 0.3 mAp-p, since the voltage drop in the feedback resistor 3 is equal to or less than 0.6 V., little current flows through the diode and the gain at transformation between an input current and an output voltage, namely, a transimpedance gain is almost determined by the feedback resistance value (2.2 k.OMEGA.) of the feedback resistor 3 and the noise characteristic is almost the same as that of a case without the diode 7. However, when the input current amplitude is over 0.3 mAp-p, the voltage drop in the feedback resistor 3 is larger than 0.6 V and the diode 7 is "on" state. At this time, the feedback resistance value Rf of the feedback resistor 3 is significantly larger than the internal resistance value Rdf of the diode and the transimpedance gain becomes almost equal to the internal resistance value Rdf of the diode (180 .OMEGA.) and the input dynamic range is enlarged to 5 mAp-p compared to about 1 mAp-p without the diode 7 (refer to FIG. 12).
FIGS. 19(a) and 19(b) are diagrams showing an input current waveform and an output voltage waveform in the preamplifier la in the above-described operation. FIG. 19(a) shows an input current waveform and an output voltage waveform when the input current amplitude is 0.2 mAp-p and FIG. 19(b) shows an input current waveform and an output voltage waveform when the input current amplitude is 2.0 mAp-p. As shown in the FIGS., while in the preamplifier 1 shown in FIG. 11 the duty cycle of the output voltage waveform varies when the input current amplitude increases to 2 mAp-p, the duty cycle does not vary even when the input current amplitude increases to 2 mAp-p in this preamplifier 1a, resulting in an output signal with no distortion.
In the prior art preamplifier a described above, a diode is connected in parallel with the feedback resistor so as to prevent a reduction in the output voltage when a large signal, namely, an input current having a large amplitude, is input, thereby enlarging the input dynamic range.
In the construction where a single diode 7 is connected in parallel with the feedback resistor 3, however, because the threshold voltage of the forward direction current of the diode connected in parallel with the feedback resistor is dependent on the material of the diode as described above, while it is possible to enlarge the dynamic range, the input signal level by which the diode is turned on, that is, an input signal level where a substantive feedback resistance value is switched, is fixed. Therefore, even if it is desirable to obtain a voltage amplitude above a certain level as the sensitivity of the amplifier when a small signal is input (for example, when a signal in a range from 0.5 to 1.0 mAp-p is input), the desired voltage value cannot be obtained when the switching point of the substantive feedback resistance value is at a signal level smaller than this input signal.
In addition, since as a characteristic of a diode, little current flows through the diode when the voltage does not exceed the Schottky barrier voltage and a current suddenly flows when the forward direction voltage exceeds the Schottky barrier voltage, that is, when the diode is turned on, the substantive feedback resistance value also varies suddenly before and after the diode is turned on. As a result, there is a problem that the linearity of the amplification is poor and a device which is connected to the amplifier at a latter stage thereof may malfunction.